Circuit for sensing an analog signal, corresponding electronic system and method

ABSTRACT

A circuit configured to sense an input analog signal generated by a sensor at a first frequency and to generate an output digital signal indicative of the sensed input analog signal. The circuit includes a conditioning circuit, an ADC, a feedback circuit, and a low-pass filter. The conditioning circuit is configured to receive the input analog signal and to generate a conditioned analog signal. The ADC is configured to provide a converted digital signal based on the conditioned analog signal. The feedback circuit includes a band-pass filter configured to selectively detect a periodic signal at a second frequency higher than the first frequency and to act on the conditioning circuit to counter variations of the periodic signal at the second frequency. The low-pass filter is configured to filter out the periodic signal from the converted digital signal to generate the output digital signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.16/738,459, filed Jan. 9, 2020, which claims priority to Italian PatentApplication No. 102019000000989, filed on Jan. 23, 2019, whichapplications are hereby incorporated herein by reference.

TECHNICAL FIELD

The present disclosure relates generally to an electronic system andmethod, and, in particular embodiments, to a circuit for sensing ananalog signal, corresponding electronic system and method.

BACKGROUND

An analog front-end (AFE) circuit generally includes analog signalconditioning circuits that use sensitive analog amplifiers, e.g.,operational amplifiers (op-amps), filters, and sometimesapplication-specific integrated circuits (ASICs) to provide anelectronic functional block which facilitates interfacing a sensor to asubsequent processing stage such as an analog-to-digital converter(ADC), a microcontroller, or the like.

Signal conditioning may be understood as manipulating an analog signalin such a way that, e.g., the signal meets the requirements of the nextstage for farther processing, e.g., in terms of maximum input (voltage)range.

In various applications, a sensing stage (e.g., a sensor) may befollowed by a signal conditioning stage (possibly involvingamplification of the signal received from the sensor) and a processingstage (comprising, e.g., an ADC and/or a microcontroller).

Operational amplifiers are generally employed to perform amplificationof the signal in the signal conditioning stage, i.e., in the AFEcircuit.

For instance, an AFE circuit may be used for reading out a signal from apiezoresistive sensor associated to a micro-electro-mechanical systems(MEMS) micro-mirror.

According to different applications, micro-mirrors may be driven in:

-   -   “linear” mode, wherein the (angular) movement of the        micro-mirror follows an approximately linear driving signal such        as a ramp signal or and saw-tooth signal; or    -   “resonant” mode, wherein the micro-mirror is driven by a        periodic driving signal (e.g., a square wave signal) at a        frequency approximately equal to the resonance frequency of the        micro-mirror itself.

In resonant mode, the frequency of the driving signal should be kept asclose as possible to the resonance frequency of the micro-mirror inorder to effectively counter damping effects and sustain vibration ofthe micro-mirror at its resonance frequency. Additionally, the drivingsignal should be synchronized with the movement of the micro-mirror,e.g., with the edges of the driving signal corresponding to zero-crossevents of a signal generated by a (piezoresistive) sensor coupled to themicro-mirror and configured to sense the position thereof.

Therefore, possible shifts of the resonance frequency of themicro-mirror during operation (e.g., due to temperature and/or pressurevariations) should be sensed in order for the driving signal to trackthe frequency shifts.

As exemplified in FIG. 1, a piezoresistive sensor 10 may be used forsensing mechanical information (e.g., deformation and/or movement) of amicro-mirror associated thereto (not visible in the Figures annexedherein).

In particular, the piezoresistive sensor 10 comprises fourpiezoresistors R1, R2, R3, and R4 in a full-bridge arrangement. Thefull-bridge arrangement is biased with a certain supply voltage betweena (positive) power supply rail BIAS_P and a (negative) power supply railBIAS_N. An analog signal V_(S) can be sensed between the intermediatenodes of the two half-bridges of the piezoresistive sensor 10, i.e.,between the node PZR_P (intermediate resistors R3 and R4) and the nodePZR_N (intermediate resistors R1 and R2).

As exemplified in FIG. 1, an analog (voltage) signal V_(S) sensedbetween the nodes PZR_P and PZR_N is propagated to a sensing circuit 12comprising an AFE circuit 121 (also referred to as conditioning circuitin the present description) configured for conditioning the analogsignal V_(S) sensed between nodes PZR_P and PZR_N.

The sensing circuit 12 may also comprise an ADC 122 configured forproviding an output digital signal ADC_out resulting from conversion todigital of the analog conditioned signal V_(S,C) provided by the AFEcircuit 121, and thus indicative of the analog signal V_(S) provided bythe piezoresistive sensor 10.

Alternatively or additionally, the sensing circuit 12 may comprise acomparator circuit 123 (e.g., a comparator with or without hysteresis)coupled at the output of the AFE circuit 121 for generating a zero-crosssignal ZC_out indicative of zero-cross events of the analog signal V_(S)provided by the piezoresistive sensor 10.

In some applications, e.g., when reading out information from resonantmicro-mirrors, the piezoresistive sensor 10 may provide asinusoidal-like analog signal V_(S) (e.g., due to periodic mechanicalmovement and/or deformation of the resonant micro-mirror) at a frequencyf_(S) corresponding to the resonance frequency of the micro-mirror. Asdiscussed previously, the resonance frequency of the micro-mirror maychange during operation, due to several root-causes related to themicro-mirror itself (e.g., variations of temperature, pressure, etc.).Therefore, a control loop (e.g., a feedback loop sensitive to signalADC_out, ZC_out or both) may be used in order to keep the micro-mirrormoving at a certain (fixed) frequency, irrespective of possiblevariations of the resonance frequency of the micro-mirror itself, and/orto adapt the frequency of the driving signal to the new (shifted)resonance frequency of the micro-mirror.

Therefore, a change of the resonance frequency of the micro-mirror maybe sensed based on the output digital signal ADC_out and/or thezero-cross signal ZC_out. However, the AFE circuit 121 itself mayintroduce absolute phase shifts and/or phase drifts in the propagatedsignal V_(S,C) with respect to the input analog signal V_(S) due to,e.g., temperature variations.

Thus, phase drift of the AFE circuit 121 should be reduced (e.g., keptas low as possible) in order not to impair a correct sensing of thedrift of the resonance frequency of the micro-mirror.

Phase drift of the AFE circuit 121 is related to the phase drift of thetransfer function V_(S,C)/V_(S) of the AFE circuit 121.

For instance, FIG. 2A is exemplary of a possible simplified transferfunction of an AFE circuit 121, in a frequency range of interest forapplications involving resonant micro-mirrors, e.g., from hundreds of Hzto tens of kHz.

In the upper portion of FIG. 2A, the magnitude G of the transferfunction V_(S,C)/V_(S) is reproduced, while in the lower portion of FIG.2A, the phase PH of the transfer function V_(S,C)/V_(S) is reproduced.

As exemplified with thick line in FIG. 2A, in the frequency range ofinterest, a simplified transfer function of an AFE circuit 121 maycomprise a single pole at a certain frequency f_(o), also referred to asthe “cut-off frequency” of the AFE circuit 121 in the presentdescription.

Generally, the AFE circuit 121 is designed so that, in normal operatingconditions, the frequency f_(o) is well above the frequency f_(S) of theanalog (sinusoidal-like) signal V_(S) received at the AFE circuit 121,thereby facilitating conditioning the input analog signal V_(S) withoutintroducing any relevant phase shift in the output analog conditionedsignal V_(S,C). For instance, the frequency f_(o) may be one decade(i.e., one order of magnitude) higher than the frequency f_(S) of theanalog signal V_(S), e.g., with f_(S)=300 Hz and f_(o)=3 kHz.

The frequency of poles and zeroes in the transfer function may vary as aresult of a change of the operating conditions of the AFE circuit 121,e.g., a change of the operating temperature.

For instance, the thin line in FIG. 2A is exemplary of a case whereinthe frequency f_(o) of the pole is lowered to a new value f_(o)′ (forinstance, in the previously considered example wherein f_(o)=3 kHz,f_(o)′ may be around 2.9 kHz). As a result, the new pole frequencyf_(o)′ may be closer than the old pole frequency f_(o) to the frequencyf_(S) of the analog signal V_(S), so that the phase of the outputconditioned signal V_(S,C) of the AFE circuit 121 may be shifted withrespect to the phase of the input signal V_(S), i.e., the outputconditioned signal V_(S,C) may exhibit a temperature-dependent shift inthe time domain, according to the following equation:

${\Delta t} = \frac{\Delta\varphi}{2\pi\; f_{s}}$

wherein Δφ is the phase shift introduced by the AFE circuit 121, f_(S)is the frequency of the input signal V_(S) and Δt is the resulting shiftin the time domain of the output conditioned signal V_(S,C).

For instance, a phase shift Δφ=10 mdeg at a signal frequency f_(S)=400Hz would result in a time shift Δt=69 ns of the output conditionedsignal V_(S,C), and thus of the output digital signal ADC_out as well asof the zero-cross assertion signal ZC_out, as exemplified in FIG. 2B. Invarious applications, such time shift Δt may not be compliant with therequirements of the control loop.

Generally, two main causes may lead to a shift of the frequency of thepoles and/or zeroes of the transfer function of an analog front-endcircuit 121.

The first main cause may be a variation of the bandwidth of one or moreoperational amplifier(s) included in the AFE circuit 121. Knownsolutions to this issue rely on designing wide-bandwidth operationalamplifiers, which however results in higher power consumption and higherdesign complexity, with corresponding increased silicon area occupationand electronic noise.

The second main cause may be a variation of the gain of the feedbacknetwork(s) included in the AFE circuit 121, e.g., due to variations ofRC time constants mainly resulting from non-negligible temperaturedependence of the resistance values. Known solutions to this issue relyon designing AFE circuits with a switched-capacitor architecture, whichhas a number of drawbacks, namely:

-   -   the need of a continuous-time antialiasing filter at the end of        the sensing chain, i.e., between the AFE circuit 121 and the ADC        122, which may still introduce temperature-related phase drifts        in the sensing chain,    -   a reduced suitability for the generation of a zero-cross signal        ZC_out due to high frequency switching,    -   a correlation between the switching frequency and the mirror        resonance frequency, resulting in intermodulation issues, and    -   an increase of silicon area and circuit complexity due to the        implementation of a plurality of programmable gain amplifiers in        the switched-capacitor architecture, e.g., due to the integrated        capacitors not being suitable for the implementation of        switched-capacitor architectures.

The piezoresistive sensor 10 may also be provided as an integratedcircuit associated to a micro-mirror. Similarly, also the sensingcircuit 12 may be provided as an integrated circuit, e.g., anapplication-specific integrated circuit. The piezoresistive sensor 10and the sensing circuit 12 may thus be mounted on a common substrate S,e.g., a printed circuit board (PCB), as exemplified in FIG. 1.

Thus, in addition to the undesired phase drift possibly generated by thetemperature-dependent behavior of the AFE circuit 121, a second sourceof possible phase drift of the analog conditioned signal V_(S,C) (andthus, of the output digital signal ADC_out and/or the zero-crossassertion signal ZC_out) is related to the temperature-dependentlow-pass filter behavior of the electrical connections on the substrateS between the piezoresistive sensor 10 and the sensing circuit 12.

In particular, parasitic capacitances C_(P) and C_(N) coupled to thenodes PZR_P and PZR_N may be present, as exemplified in FIG. 3. Suchcapacitances may comprise:

-   -   a capacitance at the pads of the micro-mirror package,    -   a capacitance due to the electrical connections between the        piezoresistive sensor 10 and the sensing circuit 12, implemented        e.g., as conductive traces on a PCB,    -   a capacitance at the pads of the package of the sensing circuit        12, and    -   an input capacitance of the AFE circuit 121.

Parasitic capacitances C_(P) and C_(N) at the output of thepiezoresistive sensor 10 combined with the piezoresistors R1, R2, R3, R4in the piezoresistive sensor 10 may result in a low-pass filterfunctionality acting on the analog signal V_(S) generated by thepiezoresistive sensor 10 (also referred to as the PZR output signal inthe present description) in a way similar to that previously describedwith reference to the transfer function of the AFE circuit 121. That is,the cut-off frequency of the low-pass filter may change due totemperature variations (e.g., due to the temperature-dependent behaviorof piezoresistors R1, R2, R3, R4), thereby generating additional phaseshifts of the analog conditioned signal V_(S,C) (and thus, of the outputdigital signal ADC_out and/or the zero-cross assertion signal ZC_out).

Known solutions to the above-mentioned issue may rely on insertingrespective compensation resistors R_(P) and R_(N) between the outputnodes of the piezoresistive sensor 10 and the input nodes of the sensingcircuit 12 as exemplified in FIG. 3, with the resistors R_(P) and R_(N)having a temperature coefficient opposite to the temperature coefficientof the piezoresistors R1, R2, R3, R4.

In particular, the piezoresistors R1, R2, R3, and R4 generally have apositive temperature coefficient (PTC), so that resistors R_(P) andR_(N) with a negative temperature coefficient (NTC) may be implementedon the PCB as exemplified in FIG. 3 in order to compensate the drift ofthe low-pass filter pole due to temperature variations.

However, sizing of the resistors R_(P) and R_(N) may be a complexprocedure, so that the solution described above allows only roughcorrections which are strongly dependent on the specific application,and a residual phase shift—oftentimes unacceptable in case of strictsystem requirements—is almost always present in the analog conditionedsignal V_(S,C).

Another known solution to the issue of temperature-dependent phase driftin AFE circuits relies on complex calibration procedures.

Calibrated AFE circuits may comprise an integrated temperature sensor,e.g., implemented as an ASIC. By calibrating the AFE circuit at two (ormore) different temperatures, an interpolation of the relationshipbetween operating temperature and phase drift of the analog conditionedsignal V_(S,C) may be inferred, so that during normal operation theoutput digital signal ADC_out may be corrected in real time based on thetemperature sensed by the integrated temperature sensor and the inferredrelationship with the phase drift of the analog conditioned signalV_(S,C).

Temperature-calibrated AFE circuits are generally expensive, and theresulting stability of the phase compensation is limited by theprecision of the calibration procedure, the accuracy of theinterpolation, and other factors. Additionally, temperature-calibratedAFE circuits require digital post-processing of the output digitalsignal ADC_out which may be often unacceptable in terms of systemrequirements.

SUMMARY

Despite the extensive activity in the area, further improved solutionsare desirable.

Some embodiments relate to a circuit, such as an AFE circuit, e.g., forsensing an analog signal from a sensor.

Some embodiments relate to a corresponding electronic system.

One or more embodiments may relate to a corresponding method ofoperating such circuit or electronic system.

One or more embodiments may be applied to AFE circuits for sensingsignals from PZR sensors. For instance, one or more embodiments may beapplied to sensing signals from piezoresistive sensors in MEMS, such asresonant micro-mirrors.

Some embodiments reduce the input-to-output phase shift of an AFEcircuit due to variations of the operating conditions (e.g., temperaturevariations) thereof, with reduced impact on silicon area and powerconsumption.

Some embodiments compensate the phase shift due to variations of theoperating conditions of the electrical connections between a sensor anda corresponding AFE circuit without the use of additional components(such as NTC resistors) coupled between the sensor and the AFE circuit.

Some embodiments provide phase shift compensation with respect tochanges of operating conditions other than temperature, e.g., variationsof the supply voltage.

Some embodiments may avoid using temperature calibration forcompensation of phase shift.

One or more embodiments relate to a circuit for sensing an input analogsignal generated by a sensor at a first frequency and generating anoutput digital signal indicative of the input analog signal sensed.

In one or more embodiments, the circuit comprises:

-   -   a conditioning circuit configured for receiving at an input port        the input analog signal and generating at an output port a        conditioned analog signal,    -   an ADC coupled at the output port of the conditioning circuit,        the ADC configured for receiving the conditioned analog signal        and providing at a converter output node a converted digital        signal resulting from conversion to digital of the conditioned        analog signal and indicative of the input analog signal sensed,    -   a feedback circuit between the converter output node of the ADC        and a control input of the conditioning circuit, the feedback        circuit comprising a band-pass filter configured to selectively        detect a periodic signal at a second frequency, the second        frequency higher than the first frequency, the feedback circuit        being configured to act on the conditioning circuit to counter        variations of the periodic signal at the second frequency, and    -   a low-pass filter coupled at the converter output node of the        ADC and having a low-pass cut-off frequency lower than the        second frequency, the low-pass filter configured to filter out        the periodic signal from the converted digital signal to        generate the output digital signal.

In one or more embodiments, the feedback circuit is configured to detectan amplitude of the periodic signal at the second frequency, and:

-   -   i) as a result of the amplitude of the periodic signal        increasing, acting on the conditioning circuit in order to lower        a cut-off frequency of the conditioning circuit; and/or    -   ii) as a result of the amplitude of the periodic signal        decreasing, acting on the conditioning circuit in order to        increase a cut-off frequency of the conditioning circuit.

In one or more embodiments, the feedback circuit is configured to detecta phase of the periodic signal at the second frequency, and:

-   -   i) as a result of the phase of the periodic signal negatively        shifting, acting on the conditioning circuit in order to        increase a cut-off frequency of the conditioning circuit; and/or    -   ii) as a result of the phase of the periodic signal positively        shifting, acting on the conditioning circuit in order to lower a        cut-off frequency of the conditioning circuit.

In one or more embodiments, the feedback circuit comprises acurrent-output digital-to-analog converter (DAC) configured to act onthe conditioning circuit to counter variations of the periodic signal atthe second frequency by varying a bias current of at least oneoperational amplifier included in the conditioning circuit.

In one or more embodiments, the feedback circuit is configured to act onthe conditioning circuit to counter variations of the periodic signal atthe second frequency by varying the value of at least one variableresistor and/or at least one variable capacitor included in theconditioning circuit.

In one or more embodiments, the feedback circuit comprises a low-passdigital filter circuit having a low-pass frequency lower than the firstfrequency and configured to filter out noise from the periodic signal atthe second frequency detected.

In one or more embodiments, the circuit further comprises a signalgenerator circuit configured for superimposing the periodic signal atthe second frequency to the input analog signal received at theconditioning circuit.

In one or more embodiments, the signal generator circuit comprises avoltage oscillator and a voltage-to-current converter circuit configuredto generate the periodic signal at the second frequency in the form of asquare wave current signal.

One or more embodiments may relate to an electronic system comprising:

-   -   a sensor biased with a supply voltage and configured to generate        a sensor analog signal at a first frequency,    -   a circuit according to one or more embodiments coupled to the        sensor for sensing the sensor analog signal at the first        frequency and configured for generating an output digital signal        indicative of the sensor analog signal sensed, and    -   a signal generator circuit configured for generating the        periodic signal at the second frequency.

In one or more embodiments the signal generator circuit is coupled tothe conditioning circuit in the circuit and is configured to superimposethe periodic signal at the second frequency to the sensor analog signalreceived at the conditioning circuit.

In one or more embodiments, the signal generator circuit is coupled tothe sensor and is configured to superimpose the periodic signal at thesecond frequency to the supply voltage which biases the sensor.

In one or more embodiments, the signal generator circuit comprises avoltage oscillator and a voltage-to-current converter circuit configuredto generate the periodic signal at the second frequency in the form of asquare wave current signal.

In one or more embodiments:

-   -   the conditioning circuit comprises a differential input stage        comprising a first operational amplifier and a second        operational amplifier each having a respective gain resistance,        and the input analog signal is received between a first input        node of the first operational amplifier and a first input node        of the second operational amplifier;    -   the signal generator circuit is configured to generate a first        periodic signal at the second frequency and a second periodic        signal at the second frequency, the second periodic signal being        in antiphase with the first periodic signal, by applying a        reference voltage to a resistance matched to the gain        resistances of the first operational amplifier and the second        operational amplifier; and    -   the first periodic signal is provided at a second input node of        the first operational amplifier, and the second periodic signal        is provided at a second input node of the second operational        amplifier.

In one or more embodiments, the sensor comprises a piezoresistive sensorcoupled to a micro-mirror and configured to detect motion of themicro-mirror.

One or more embodiments, relate to a method of operating a circuit or anelectronic system, according to one or more embodiments, the methodcomprising:

-   -   receiving at the input port of the conditioning circuit an input        analog signal generated by a sensor at a first frequency and        conditioning the input analog signal to generate at an output        port of the conditioning circuit a conditioned analog signal;    -   receiving the conditioned analog signal at the ADC coupled at        the output port of the conditioning circuit and converting the        conditioned analog signal to a converted digital signal        indicative of the input analog signal sensed;    -   selectively detecting a periodic signal at a second frequency        higher than the first frequency at the feedback circuit and        acting on the conditioning circuit to counter variations of the        periodic signal at the second frequency; and    -   filtering out the periodic signal at the second frequency from        the converted digital signal to generate the output digital        signal.

BRIEF DESCRIPTION OF THE DRAWINGS

One or more embodiments will now be described, by way of example only,with reference to the annexed figures, wherein:

FIG. 1 shows an exemplary piezoresistive sensor and sensing circuit;

FIG. 2A shows an example of a possible simplified transfer function ofthe AFE of FIG. 1.

FIG. 2B shows an example of possible waveforms of the sensing circuit ofFIG. 1;

FIG. 3 illustrates the presence of parasitic capacitances in the circuitof FIG. 1;

FIG. 4 is a circuit diagram exemplary of a possible architecture of anAFE circuit;

FIG. 5 is a circuit diagram exemplary of embodiments;

FIG. 6 is exemplary of implementation details of the embodiments of FIG.5;

FIG. 7 is a circuit diagram exemplary of further embodiments; and

FIG. 8 is exemplary of implementation details of the embodiments of FIG.7.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

In the ensuing description, one or more specific details areillustrated, aimed at providing an in-depth understanding of examples ofembodiments of this description. The embodiments may be obtained withoutone or more of the specific details, or with other methods, components,materials, etc. In other cases, known structures, materials, oroperations are not illustrated or described in detail so that certainaspects of embodiments will not be obscured.

Reference to “an embodiment” or “one embodiment” in the framework of thepresent description is intended to indicate that a particularconfiguration, structure, or characteristic described in relation to theembodiment is comprised in at least one embodiment. Hence, phrases suchas “in an embodiment” or “in one embodiment” that may be present in oneor more points of the present description do not necessarily refer toone and the same embodiment. Moreover, particular conformations,structures, or characteristics may be combined in any adequate way inone or more embodiments.

Throughout the figures annexed herein, like parts or elements areindicated with like references/numerals and a corresponding descriptionwill not be repeated for brevity.

The references used herein are provided merely for convenience and hencedo not define the extent of protection or the scope of the embodiments.

By way of general introduction to the detailed description of exemplaryembodiments, reference may be first had to FIG. 4.

FIG. 4 is a circuit diagram exemplary of a possible architecture of anAFE circuit 121, having input nodes PZR_P and PZR_N for coupling to theoutput of a sensor 10 (e.g., a piezoresistive sensor cooperating with amicro-mirror) and output nodes AFE_P and AFE_N for providing aconditioned replica V_(S,C) of the input signal V_(S) received from thesensor 10.

In particular, an AFE circuit 121 comprises analog amplifiers such asoperational amplifiers which implement a single-stage or multi-stageamplifier circuit. Various stages of the AFE circuit 121 are coupled byusing RC networks, optionally tunable (e.g., by using variableresistors, as exemplified in FIG. 4, and/or variable capacitors). Theinput stage of the AFE circuit 121 may comprise a pair of operationalamplifiers, having respective first input nodes coupled to the nodesPZR_P and PZR_N for receiving the input signal V_(S).

Generally, the frequency behavior of an AFE circuit 121 as exemplifiedin FIG. 4 may be approximated, in the frequency range of interest formicro-mirror applications, with a single-pole transfer function asalready discussed with reference to FIG. 2A. Generally, the frequencyf_(o) of such single pole may be a decade higher than the frequencyf_(S) of the input signal V_(S) received at nodes PZR_P and PZR_N. Forexample, f_(o) may have a value in the range from a few kHz (e.g., 3kHz) to hundreds of kHz (e.g., 700 kHz).

The transfer function of an AFE circuit 121 may comprise further polesand/or zeroes, which are however typically located at frequencies whichare not of interest in the context of the present description.

According to some embodiments, e.g., and as exemplified in FIG. 5, acircuit 12 for sensing an analog signal V_(S) generated by a sensor 10comprises:

-   -   an AFE circuit 121 configured for conditioning an analog input        signal V_(S) sensed between nodes PZR_P and PZR_N, thereby        generating an analog conditioned signal V_(S,C);    -   an ADC 122 configured for providing a digital output signal        ADC_out resulting from conversion to digital of the analog        conditioned signal V_(S,C) provided by the AFE circuit 121;    -   a feedback circuit 125 coupled between the output node of the        ADC 122 and a control input of the AFE circuit 121, the feedback        circuit 125 being configured to detect, at the output of the ADC        122, a reference periodic signal at a frequency f_(H) higher        than (or comparable to) the cut-off frequency f_(o) of the AFE        circuit 121 (thus, also higher than the frequency f_(S):        f_(H)>f_(o)>f_(S)) propagated through the AFE circuit 121 and        the ADC 122, and act on the AFE circuit 121 to counter amplitude        and/or phase variations of the reference periodic signal        detected; and    -   a digital low-pass filter 126 coupled at the output of the ADC        122, the digital low-pass filter 126 having a cut-off frequency        f₁₂₆ lower than the cut-off frequency f_(o) of the AFE circuit        121 and lower than the frequency f_(H) of the reference periodic        signal.

Therefore, the digital low-pass filter 126 is configured to filter thedigital output signal ADC_out to provide a further digital output signalADC_out′ at frequency f_(S) indicative of the analog signal V_(S)generated by the sensor 10, and not disturbed by the propagation of asignal at frequency f_(H) through the AFE circuit 121 and the ADC 122.

The frequency f_(H) is preferably higher than f_(o) (e.g.,f_(H)>2f_(o)), e.g., to facilitate filtering out the reference periodicsignal at the low-pass filter 126.

However, in some embodiments, operation of a circuit as disclosed hereinmay be effective provided that the frequency f_(H) is “comparable” tothe cut-off frequency f_(o), i.e., provided that propagation of a signalat frequency f_(H) through the AFE circuit 121 is affected (in terms ofamplitude and/or phase) by variations (even slight variations) of thecut-off frequency f_(o). To this regard, frequency f_(H) may also belower than the cut-off frequency f_(o).

In one or more embodiments, the reference periodic signal at frequencyf_(H) is superimposed to the input signal V_(S) received at nodes PZR_Pand PZR_N of the AFE circuit 121.

In a circuit arrangement as described above and illustrated in FIG. 6,the reference periodic signal at frequency f_(H) (e.g., a square wavesignal) propagates through the AFE circuit 121 to the output nodes AFE_Pand AFE_N, and thus through the ADC circuit 122, in a way analogous tothat of the input analog signal V_(S) received at nodes PZR_P and PZR_Nfrom the sensor 10 (also referred to as the “effective input signal” inthe present description).

As a result of the frequency f_(H) being higher than the frequency f_(S)and higher than (or comparable to) the cut-off frequency f_(o),propagation (in terms of amplitude and/or phase) of the referenceperiodic signal at frequency f_(H) is affected (also) by slightvariations of the cut-off frequency f_(o) (e.g., due to temperaturevariations).

For instance, as a result of the cut-off frequency f_(o) (slightly)decreasing, the amplitude of the reference periodic signal propagated atthe output of the ADC circuit 122 may decrease (i.e., attenuation maytake place) and the phase may negatively shift.

Conversely, as a result of the cut-off frequency f_(o) (slightly)increasing, the amplitude of the reference periodic signal propagatedmay increase (i.e., amplification may take place) and the phase maypositively shift.

Therefore, sensing amplitude and/or phase variations of the referenceperiodic signal at frequency f_(H) may facilitate acting on the AFEcircuit 121 in order to compensate (even slight) variations of thecut-off frequency f_(o), with the aim of letting the effective inputsignal V_(S) at frequency f_(S) propagate unaffected.

The feedback circuit 125 is configured to filter the output digitalsignal ADC_out sensed at the output of the ADC 122 with a band-passfilter 1251 centered around the frequency f_(H) of the referenceperiodic signal (i.e., f₁₂₅₁≈f_(H)).

The feedback circuit 125 may thus be configured to measure the amplitudeof such filtered signal at the frequency f_(H), e.g., measuring theamplitude of the 1st harmonic, and to drive a current-outputdigital-to-analog converter (DAC) 1253 in order to vary the bias currentof at least one operational amplifier in the AFE circuit 121 so as tocounter (any) variation of the amplitude of the filtered signal, withthis operation also resulting in countering (any) variation (or shift)of the frequency f_(o) of the main pole of the AFE circuit 121.

Alternatively or additionally, in some embodiments, the feedback circuit125 may be configured to measure the phase of such filtered signal atthe frequency f_(H) and to drive the current-output DAC converter 1253in order to vary the bias current of at least one operational amplifierin the AFE circuit 121 so as to counter (any) variation of the phase ofthe filtered signal, with this operation also resulting in counteringany variation (or shift) of the frequency f_(o) of the main pole of theAFE circuit 121.

Alternatively or additionally, the feedback circuit 125 may beconfigured to operate on parameter(s) of at least one operationalamplifier in the AFE circuit 121 other than the bias current, in orderto counter (any) variation (be it an amplitude variation or a phasevariation) of the filtered signal at the frequency f_(H).

Alternatively or additionally, in some embodiments, the feedback circuit125 may be configured to operate on other parameters of the AFE circuit121, even not related to the operational amplifiers included therein(e.g., values of resistors and/or capacitors).

Therefore, the feedback circuit 125 implements a sort of control loopoperating in parallel with the normal behavior of the AFE circuit 121,in order to (indirectly) measure the sign and magnitude of the phaseshift of the AFE circuit 121 (e.g., due to temperature variations) andconsequently adjust the biasing and/or other parameters of the AFEcircuit 121 to compensate such phase shift.

As described, such measuring the sign and magnitude of the phase shiftof the AFE circuit 121 may be performed by:

-   -   directly measuring the phase shift of the reference periodic        signal propagated through the AFE circuit 121 and the ADC        circuit 122, and/or    -   measuring the amplitude variation of the reference periodic        signal propagated through the AFE circuit 121 and the ADC        circuit 122, and correlating such amplitude variation to a        corresponding phase shift.

In the above exemplified embodiments, the feedback circuit 125 takes asdigital input the signal ADC_out and operates through discrete steps inorder to keep constant the amplitude and/or the phase of the propagatedreference periodic signal, thereby facilitating keeping constant thecut-off frequency f_(o) of the main pole of the transfer function of theAFE circuit 121.

The width of such discrete steps should be lower than the maximum phaseshift that can be tolerated in the specific application, and the numberof available steps should be related to the range of the phase shift forwhich compensation is sought.

Optionally, in some embodiments, additional digital filtering may beimplemented in the feedback circuit 125 by a digital filter circuit 1252between the band-pass filter 1251 and the current-output DAC converter1253, e.g., to filter out noise and provide a low-frequency feedbackbehavior of the feedback circuit 125. The amplitude of the signal at thefrequency f_(H) filtered by the band-pass filter 1251 may be quite lowand possibly may have a low signal-to-noise ratio (SNR), so thatlow-pass digital filtering at the filter circuit 1252 may facilitatedetecting slow variations of the feedback signal (e.g., over a timescale comparable with the typical time scale of temperature variations)and thus provide an improved dynamic range of the measured feedbacksignal.

Therefore, in some embodiments, additional digital filtering performedat the digital filter circuit 1252 may be advantageous in providing alow-frequency compensation signal (e.g., a current compensation signalwith frequency lower than f_(S)) to improve stability of the system.

In one or more embodiments, the circuit 12 may comprise a signalgenerator circuit 124, configured to generate the reference periodicsignal at the frequency f_(H) and providing such reference periodicsignal to the AFE circuit 121 so to be superimposed to the analog inputsignal V_(S).

In particular, as exemplified in FIG. 6, the signal generator circuit124 is configured for generating a reference periodic signal such as asquare wave signal, a sinusoidal signal, or others.

For instance, the signal generator circuit 124 may comprise a voltageoscillator and a voltage-to-current (V2I) circuit in order to generate asquare-wave reference current signal. In some embodiments, generatingthe reference periodic signal(s) by using a voltage oscillator and avoltage-to-current circuit may facilitate reducing the dependency on thegain resistance of the AFE circuit 121.

As exemplified in FIG. 6, the reference periodic signal(s) may be inputas current signal(s) at the inverting inputs of the operationalamplifiers in the first stage of the AFE circuit 121. Such currentsignals result in an equivalent input voltage having a value whichdepends on the current magnitude and on the gain resistance of the firststage of the AFE circuit 121, and which is further amplified in thefollowing stages of the AFE circuit 121. The reference periodic currentsignals may be generated by applying a precise andtemperature-independent reference voltage (e.g., a bandgap reference) toa resistance of the same technological type as the gain resistance andmatched thereto, so that the magnitude of the equivalent input voltagemay not depend on the value of the reference periodic current signal(s)or the value of the gain resistance. In some embodiments, the magnitudeof the equivalent input voltage may depend (almost) only on the value ofthe reference voltage, so to be in turn almost temperature-independent.

Optionally, in some embodiments, the signal generator circuit 124 maygenerate a pair of reference periodic signals, e.g., square-wave currentsignals, with the second signal in the pair being anti-phased withrespect to the first signal in the pair.

The first signal in the pair of reference periodic signals may beprovided at a second input of the first operational amplifier of theinput stage of the AFE circuit 121 (with the first input of the firstoperational amplifier configured for coupling to node PZR_P), and thesecond signal in the pair of reference periodic signals may be providedat a second input of the second operational amplifier of the input stageof the AFE circuit 121 (with the first input of the second operationalamplifier configured for coupling to node PZR_N).

It will be understood that providing a pair of anti-phased referenceperiodic signals as exemplified in FIG. 6 may be advantageous in thecontext of a fully-differential architecture as illustrated. However, inone or more embodiments a single reference periodic signal may beprovided to the AFE circuit 121.

According to one or more embodiments, e.g., and as exemplified in FIG.7, the signal generator circuit 124 may be configured for providing thereference periodic signal at a frequency f_(H) to the sensor 10 insteadof the AFE circuit 121.

In the embodiments of FIG. 7, the reference periodic signal issuperimposed to the “effective input signal” V_(S) generated by thesensor 10 directly at the sensor 10.

For instance, as exemplified in FIG. 8, the reference periodic signalmay be superimposed to the bias voltage of the piezoresistive sensor 10,being propagated to the output signal V_(S) thereby.

Therefore, in the circuit arrangement illustrated in FIG. 8, thereference periodic signal at frequency f_(H) propagates not only throughthe AFE circuit 121, but also through the electrically conductive lineswhich couple the sensor 10 to the sensing circuit 12.

As a result, propagation (in terms of amplitude and/or phase) of thereference periodic signal at frequency f_(H) may be affected not only byvariations of the cut-off frequency f_(o), but also by variations of thefrequency of the low-pass filter pole due to the parasitic capacitancesC_(P) and C_(N) combined with the piezoresistors R1, R2, R3, and R4 inthe piezoresistive sensor 10.

Embodiments according to the topology illustrated in FIGS. 7 and 8,therefore, advantageously provide means for compensating phase shifts ofa whole “sensor plus circuit” system, by making the reference periodicsignal propagate through the entire propagation path of the effectivesignal V_(S).

It will be appreciated that various additional or optional featuresdescribed with reference to the embodiments of FIGS. 5 and 6 may applyalso to the embodiments of FIGS. 7 and 8.

Embodiments of the present disclosure thus may facilitate reducing phaseshift phenomena in AFE circuits, e.g., due to temperature variations, byusing a feedback loop configured for sensing a reference periodic signalsuperimposed to the effective input signal and having a frequency higherthan the frequency of the effective input signal, with improvedperformance over the prior art approaches.

Additionally, embodiments of the present disclosure may provide thepossibility of implementing AFE circuits, such as 121, with an“aggressive” low-pass filter functionality (i.e., with a dominant polefrequency f_(o) closer to the signal frequency f_(S) if compared toprior art solutions) which may advantageously provide improved noiserejection performance.

One or more embodiments are suitable for use with continuous-time AFEcircuits and rely on few additional circuits for implementing phaseshift compensation, thereby resulting in a negligible increase ofsilicon area and power consumption if compared to non-compensatedcircuits.

One or more embodiments may facilitate compensating the effect ofnon-negligible parasitic capacitances C_(P), C_(N) and/or the effect ofpiezoresistors with PTC behavior without the need of NTC resistors beingmounted on the PCB.

One or more embodiments are effective in providing phase shiftcompensation not only with respect to temperature variations, but alsowith respect to power supply variations or other variations of operatingconditions.

Embodiments of the present description advantageously do not require anexpensive temperature calibration process.

Without prejudice to the underlying principles, the details andembodiments may vary, even significantly, with respect to what has beendescribed byway of example only, without departing from the extent ofprotection.

What is claimed is:
 1. A sensing circuit comprising: a conditioningcircuit configured to receive, at an input port, an input analog signalgenerated by a sensor at a first frequency, and to generate, at anoutput port, a conditioned analog signal; an analog-to-digital converter(ADC) coupled to the output port of the conditioning circuit, the ADCconfigured to receive the conditioned analog signal and provide, at aconverter output node, a converted digital signal based on theconditioned analog signal, the converted digital signal being indicativeof the input analog signal; a feedback circuit coupled between theoutput port of the conditioning circuit and a control input of theconditioning circuit, the feedback circuit comprising a band-pass filterconfigured to selectively detect a periodic signal at a secondfrequency, the second frequency being higher than the first frequency,wherein the feedback circuit is configured to act on the conditioningcircuit to counter variations of the periodic signal at the secondfrequency; and a signal generator circuit configured to superimpose theperiodic signal at the second frequency to the input analog signalreceived at the conditioning circuit.
 2. The sensing circuit of claim 1,wherein the feedback circuit is coupled to the output port of theconditioning circuit via the ADC.
 3. The sensing circuit of claim 1,further comprising a low-pass filter coupled to the converter outputnode of the ADC and having a low-pass cut-off frequency higher than thefirst frequency, the low-pass filter configured to filter out theperiodic signal from the converted digital signal to generate an outputdigital signal indicative of the input analog signal.
 4. The sensingcircuit of claim 3, wherein the low-pass cut-off frequency is lower thanthe second frequency.
 5. The sensing circuit of claim 3, wherein thelow-pass cut-off frequency is higher than the second frequency.
 6. Thesensing circuit of claim 1, wherein the periodic signal is a square wavesignal or a sinusoidal signal.
 7. The sensing circuit of claim 1,wherein the periodic signal is a current signal.
 8. The sensing circuitof claim 1, wherein the feedback circuit is configured to detect anamplitude or phase of the periodic signal at the second frequency andact on the conditioning circuit based on the detected amplitude or phaseto counter variations of the periodic signal at the second frequency. 9.The sensing circuit of claim 1, wherein the signal generator circuitcomprises a voltage oscillator and a voltage-to-current convertercircuit configured to generate the periodic signal at the secondfrequency in the form of a square wave current signal.
 10. The sensingcircuit of claim 1, wherein the signal generator circuit is configuredto superimpose the periodic signal at the second frequency to the inputanalog signal received at the conditioning circuit by injecting theperiodic signal into a supply voltage of the sensor.
 11. The sensingcircuit of claim 1, wherein the conditioning circuit comprises anoperational amplifier, and wherein the signal generator circuit isconfigured to superimpose the periodic signal at the second frequency tothe input analog signal received at the conditioning circuit byinjecting the periodic signal directly into an input of the operationalamplifier.
 12. The sensing circuit of claim 11, wherein the input analogsignal is a differential signal, and wherein the periodic signal is adifferential signal.
 13. The sensing circuit of claim 1, wherein thesignal generator circuit is configured to generate the periodic signalby applying a temperature-independent reference voltage to a resistor.14. A micro-electro-mechanical (MEM) system comprising: a MEM actuator;a piezoresistive sensor configured to sense mechanical information ofthe MEM actuator and produce an input analog signal based on themechanical information of the MEM actuator; and a sensing circuitcomprising: a conditioning circuit configured to receive, at an inputport, the input analog signal, and to generate, at an output port, aconditioned analog signal; an analog-to-digital converter (ADC) coupledto the output port of the conditioning circuit, the ADC configured toreceive the conditioned analog signal and provide, at a converter outputnode, a converted digital signal based on the conditioned analog signal,the converted digital signal being indicative of the input analogsignal; a signal generator circuit configured to superimpose a periodicsignal to the input analog signal; and a feedback circuit coupledbetween the output port of the conditioning circuit and a control inputof the conditioning circuit, wherein the feedback circuit is configuredto act on the conditioning circuit to counter variations of the periodicsignal.
 15. The MEM system of claim 14, wherein the signal generatorcircuit is configured to inject the periodic signal to a supply node ofthe piezoresistive sensor.
 16. The MEM system of claim 14, wherein theconditioning circuit comprises an operational amplifier configured toreceive the analog input signal, and wherein the signal generatorcircuit is configured to inject the periodic signal directly to an inputof the operational amplifier.
 17. The MEM system of claim 14, whereinthe MEM actuator is a micro-mirror and wherein the mechanicalinformation comprises deformation or movement of the micro-mirror.
 18. Amethod comprising: generating, with a sensor, an input analog signal ata first frequency; superimposing a periodic signal at a second frequencyto the input analog signal to generate a combined analog signal, whereinthe second frequency is higher than the first frequency; generating,using a conditioning circuit, a conditioned analog signal based on thecombined analog signal; selectively detecting the periodic signal fromthe conditioned analog signal; and acting on the conditioning circuit tocounter variations of the periodic signal detected from the conditionedanalog signal.
 19. The method of claim 18, further comprising:converting the conditioned analog signal to a converted digital signal;and filtering out the periodic signal from the converted digital signalto generate an output digital signal, wherein selectively detecting theperiodic signal from the conditioned analog signal comprises selectivelydetecting the periodic signal from the converted digital signal.
 20. Themethod of claim 18, wherein superimposing the periodic signal to theinput analog signal to generate the combined analog signal comprisesinjecting the periodic signal to a supply terminal of the sensor. 21.The method of claim 18, wherein superimposing the periodic signal to theinput analog signal to generate the combined analog signal comprisesinjecting the periodic signal to an operational amplifier of theconditioning circuit.
 22. The method of claim 18, further comprisinggenerating the periodic signal by applying a temperature independentreference voltage to a resistor.